Technical Field
The present disclosure relates to control devices for quasi-resonant switching converters; and further to corresponding control methods.
Description of the Related Art
Power switching converters (also called “switching regulators”) are known, which are designed to convert a quantity received at input, for example an AC voltage coming from the electrical network, into a regulated output quantity, for example a DC voltage.
Such converters are generally required to meet stringent requirements as regards the corresponding electrical performance, for example, to guarantee a high quality factor, or a substantially unitary power factor.
A control mode that has proven effective is the quasi-resonant mode; FIG. 1 illustrates by way of example the configuration of a flyback converter. It is emphasized, however, that what follows also applies to different types of converters, for example those of a buck-boost type.
The converter, designated as a whole by 1, comprises a transformer 2, having a primary winding 2a, a secondary winding 2b, and an auxiliary winding 2c. 
The primary winding 2a has a first terminal 2a′ connected to a supply line 3, for example to the electrical mains supplying an AC line voltage VAC, through a rectifier stage 4 that provides an input voltage Vin, and a second terminal 2a″ connected to a switch element 5, for example a MOSFET.
The switch element 5 has a first current-conduction terminal, in particular the drain terminal of the respective MOSFET, connected to the aforesaid second terminal 2a″ of the primary winding 2a, and a second current-conduction terminal, in particular the source terminal of the respective MOSFET, connected to a first reference terminal (ground, GND), through a detection resistor 6.
The switch element 5 and the detection resistor 6 define between them a first feedback node FB1, providing a first feedback voltage VCS, which is a function of the current flowing through the primary winding of the transformer 2.
The secondary winding 2b has a respective first terminal 2b′ connected to a first output terminal Out1, via a diode element 7 (having its anode connected to the same first terminal 2b′ and its cathode connected to the first output terminal Out1), and a respective second terminal 2b″ connected to a second output terminal Out2. A charge-storage element 8 is connected between the first and second output terminals Out1, Out2, in particular a capacitor, on which an output voltage Vout is present, for example a DC voltage.
The auxiliary winding 2c has a respective first terminal 2c′ and a respective second terminal 2c″ connected to a resistive divider formed by a first division resistor 9a and by a second division resistor 9b, defining between them a second feedback node FB2, on which a second feedback voltage VZCD is present.
The converter 1 further comprises a control device 10 (also defined as “controller”), which, on the basis of the first and second feedback voltages VCS, VZCD, received on respective input pins, controls in pulse-width modulation (PWM) opening and closing of the switch element 5, via a control signal Sc provided to the gate terminal of the corresponding MOSFET.
In detail, the control device 10 implements management of the switch element 5 in a quasi-resonant mode with peak-current control, which envisages two distinct phases that follow one another cyclically:
1) an energy-storage phase, during which the switch element 5 is closed (the corresponding MOSFET is on, ‘ON’ interval of the duty cycle) so as to store energy in the primary winding 2a of the transformer 2, with the diode element 7 preventing the current in the secondary winding 2b from reaching an output load (here not represented). This step terminates (triggering the subsequent energy-transfer step) when the first feedback voltage VCS reaches a threshold defined by a closed control loop (based upon a peak-current control); and
2) an energy-transfer phase, during which the switch element 5 is open (the corresponding MOSFET is off, ‘OFF’ interval of the duty cycle), so as to transfer the energy previously stored in the primary winding 2a of the transformer 2 to the secondary winding 2b and the load connected at the output. Completion of energy transfer is signaled by onset of a condition of resonance on the primary of the transformer 2, on account of the capacitance present on the drain terminal of the MOSFET of the switch element 5. This phase terminates (once again triggering the energy-storage phase) when the second feedback voltage VZCD drops below a lower threshold close to zero. This control is defined as “zero-current detection” (ZCD) control.
In greater detail, and as illustrated in FIG. 2, closing of the switch element 5 (determined by the control signal Sc, which is also represented in FIG. 2) is based upon a peak-current control mode. The current that circulates in the primary of the transformer 2 (designated by IP in FIG. 2) is compared with a sinusoidal reference current, in phase with the line voltage VAC, generated by the closed control loop for determining the instant of opening of the switch element 5 (and of turning-off of the corresponding MOSFET).
The envelope of the peaks IPK of the primary current IP has a sinusoidal waveform, whereas the current effectively absorbed by the line, designated by IL, represents the mean value of the same primary current IP. This current IL is practically sinusoidal and in phase with the line voltage VAC, thus enabling a desired correction of the power factor.
In order to implement the quasi-resonant control mode, the switch element 5 is closed (and the corresponding MOSFET is turned on) at a minimum of the resonant oscillation present on the drain voltage of the corresponding MOSFET, when the transformer 2 completes energy transfer to the secondary winding (reaching a demagnetization condition). It has indeed been shown that the switching losses are markedly reduced if turning-on of the MOSFET occurs when the drain voltage is minimum or close to zero.
FIG. 3 shows the drain voltage, the gate voltage, coinciding with the control signal Sc, and also the second feedback voltage VZCD. In order to highlight the oscillation, the figure shows the waveforms that these voltages would assume, in the case where the switch element 5 were not closed again in order to implement the quasi-resonant operation described previously.
As highlighted in FIG. 3, the drain voltage, upon turning-off of the MOSFET (upon opening of the switch element 5), increases from a substantially zero value up to a value substantially equal to the sum of the input voltage Vin and a voltage VR, which corresponds to the output voltage Vout fed back onto the primary (i.e., multiplied by the ratio of the turns between the primary and secondary windings 2a, 2b of the transformer 2), which it reaches after a settling interval during which oscillations due to the leakage inductances of the transformer 2 occur.
Next, when the energy transfer is completed, the drain voltage starts to oscillate in a resonance condition, with an amplitude of the oscillation equal to Vin+VR, with a mean value equal to Vin.
To establish the instant of turning-on of the MOSFET, the control device 10 uses the second feedback voltage VZCD, which is a function of the auxiliary voltage Vaux. When the current on the secondary of the transformer 2 goes to zero, the voltage on the diode element 7 is zero, so that the voltage on the secondary winding 2b (and consequently an auxiliary voltage Vaux across auxiliary winding 2c) is proportional to the output voltage Vout.
The control device 10 is thus configured for detection of the “valleys” of the second feedback voltage VZCD, when, that is, the second feedback voltage VZCD drops below a lower threshold, or reaches a substantially zero value.
In detail, with reference to FIG. 4, the control device 10 is configured to analyze the plot of the second feedback voltage VZCD, obtained starting from the aforesaid auxiliary voltage Vaux by the resistive divider formed by the division resistors 9a, 9b. 
The control device 10 compares, in a comparator, the value of the second feedback voltage VZCD with a first threshold Th1, referred to as an “arming threshold”. When the second feedback voltage VZCD exceeds the first threshold Th1, an arming signal ARM is switched, for example to the high logic value, and the comparator is enabled for a subsequent comparison between the same second feedback voltage VZCD and a second threshold Th2, referred to as “trigger threshold”, of a value lower than the first threshold Th1 and close to zero.
When the second feedback voltage VZCD drops below the aforesaid second threshold Th2, a trigger signal TRIG is switched, for example to the high logic value, and the control device 10 detects a condition indicating occurrence of a valley of the auxiliary voltage Vaux, and thus indicating that demagnetization has occurred, thus determining closing of the switch element 5.
Crossings of the second threshold Th2 that occur during a blanking interval, designated by Tblank, of a preset minimum value starting from opening of the switch element 5, are not considered, in order to prevent spurious oscillations on the auxiliary voltage Vaux from possibly causing false detections.
In FIG. 4, detection of the valley, which causes closing again of the switch element 5 (and the end of the ‘OFF’ interval of the PWM control signal Se), occurs after a detection time interval, designated by TZCD, starting from the previous opening of the switch element 5.
Even though the converter 1 has generally good electrical performance, the performance is not optimized, at least as regards certain operating conditions.